Design Guidelines for RT7302 and RT7304 PSR LED Driver
Abstract
RT7302 and RT7304 are constant current LED drivers with active power factor correction (PFC). They support high power factor across a wide range of line voltages, and drive the converter in the quasiresonant (QR) mode to achieve higher efficiency. By using primary side regulation (PSR), RT7302/RT7304 control the output current accurately without the need of a shunt regulator or optocoupler at the secondary side, reducing the external component count, the cost, and the size of the driver board.
This application note presents a step by step design guideline for an isolated single stage constant current LED driver with PFC using the RT7302. The guideline can also be applied to RT7304.
The design example in this application note describes an 18W LED driver with slim formfactor, suitable for T8 LED tube applications, but the same design can also be used in LED bulb or other form factor applications.
1. Introduction
RT7302 and RT7304 are constant current LED drivers with active power factor correction (PFC). They support high power factor across a wide range of line voltages, and drive the converter in the quasiresonant (QR) mode to achieve higher efficiency. By using primary side regulation (PSR), RT7302/RT7304 control the output current accurately without a shunt regulator or optocoupler at the secondary side, reducing the external component count, the cost, and the size of the LED driver board.
RT7304 embeds comprehensive protection functions for robust designs, including LED open circuit protection, LED short circuit protection, output diode short circuit protection, VDD under voltage lockout (UVLO), VDD overvoltage protection (VDD OVP), overtemperature protection (OTP), and cyclebycycle current limitation. RT7304 is available in a cost effective SOT236 package.
RT7302 has the same basic functionality as RT7304, but integrates more features, including fast startup via high voltage pin, PWM dimming, and input voltage feedforward compensation. RT7302 is available with SOP8 package.
This application note presents a step by step design guideline for an isolated single stage constant current LED driver with PFC using the RT7302. The design guideline can also be applied to RT7304.
The design example in this application note is an 18W LED driver with slim formfactor, suitable for T8 LED tube applications.
Figure 1. Picture of the 18W evaluation board with a typical T8 LED assembly
2. RT7302 Basic Operation
Figure 2 shows RT7302 in a typical flyback converter topology with input voltage (V_{in}).
Figure 2
When main switch Q1 is turned on with a fixed ontime t_{on}, the peak current I_{L_pk} of the magnetic inductor L_{m} can be calculated by the following equation:
If the input voltage is the output voltage of the fullbridge rectifier with sinusoidal input voltage V_{in_pk}·sin(θ), the inductor peak current I_{L_pk} can be expressed as the following equation:
When the converter operates in criticalconduction mode (CRM) with constant ontime control, the envelope of the peak inductor current will follow the input voltage waveform inphase. Thus, high power factor can be achieved. The minimum on time is set by the upper divider resistor of the ZCD network R_{ZCD1}.
Quasi resonant switching is achieved by sensing the auxiliary winding zero current condition and a smart internal valley detection circuit. Switchon of the MOSFET will always happen at a valley of the resonant voltage, thereby reducing switching losses and EMI. The ZCD pin is also used to sense output OVP condition, set by R_{ZCD2}.
The primary peak current is sensed by measuring the voltage across the MOSFET source resistor via the CS pin. An internal leading edge blanking circuit removes any spikes from this signal. A current variation due to a propagation delay is compensated by the CS pin internal current source and the external series resistor R_{PC}.
The MULT pin is used for sensing the input peak voltage, and controls the ramp for Ton generation. The line voltage sense is used as feedforward to adjust the ramp for constant COMP voltage over line voltage. This improves the regulation, eases compensation and achieves accurate maximum power limit over the full mains range. This is especially important for full range LED driver designs.
RT7302 HV pin will quickly charge the capacitor connected to the VDD pin during startup. After startup, the HV pin is disconnected, and the VDD is supplied by the auxiliary winding. This method ensures fast startup without extra power dissipation in the bleeder resistor during normal operation.
Design Procedure:
The basic design sequence is as following:
Define Input and output conditions → Calculate input power → Transformer design, Calculate N ratio, Primary inductance, Primary and secondary winding turns → Current sense resistor (R_{CS}), bridge rectifier, MOSFET parameters, Output diode parameters → Minimum ton setting (R_{ZCD1}) → OVP setting (R_{ZCD2}) → Propagation delay setting (R_{PC}) → Feedforward compensation (R_{M1}, R_{M2})
The RT7302 design tool can be used to quickly determine the component values. Chapter 3 contains a detailed step by step design description for the 18W reference design.
3. Design of an 18W LED Driver for T8 Applications
The LED driver example for this section is the 18W T8 LED driver evaluation board, see Figure 3.
Figure 3. The board measures 230x18x10mm and will fit in most narrow tube T8 housing behind the LED board.
Requirement specification for this design:
 Input range:
90V ~ 264V_{ac}
 LED load: 45V /
400mA
 Efficiency
>85% at 120V / 230V_{ac}
 PF: > 0.95
and THDi < 15% (meet IEC6100032 class C & D)
Step 1. Input and Output Conditions
The input and output conditions are listed as follows:
maximum AC input voltage V_{ac_max}, 264
V_{ac}
minimum AC input voltage V_{ac_min}, 90V_{ac}
line frequency f_{line}, 50Hz / 60Hz
average output current I_{O}, 400mA
minimum average output voltage V_{o_min}, 43V
maximum average output voltage V_{o_max}, 47V
LED string is using 14 high power LEDs with total dynamic resistance of 14Ω
Estimated maximum average input power P_{in_max_est} can be expressed as:
where η is the estimated efficiency.
The efficiency is estimated at 85%, the input power will become : 47*0.4/0.85 = 22.12W.
Estimated peak current transfer ratio of the transformer (CTR_{TX1}) can be expressed as
where I_{SEC_pk} is the peak current of secondary side, I_{PRI_pk} is the peak current of the primary side, N_{S} is the turn’s number of the secondary winding, and N_{P} is the turn’s number of the primary winding. CTR_{TX1} can be estimated to be 0.9.
The reflected output voltage Vro can be express as
where V_{f} is the forward voltage of output diode. V_{ro} is recommended to be within 95 ~125V.
In the example: Set V_{ro} = 125V.
Min. VDD supply voltage at max. output voltage V_{DD_Vomax_min} can be derived as
where V_{TH_OFF} is the falling under voltage lockout (UVLO) threshold voltage of the controller.
VDD supply voltage at max. output voltage V_{DD_max} must be within V_{DD_Vomax_min} ~ V_{DD_OVP_min}.
In the example:
V_{O_max }= 47V, V_{O_min }= 43V, V_{TH_OFF_max }= 10V, V_{DD_Vomax_min}= 14.2V
Set V_{DD_max }= 20V.
Output Capacitor C_{OUT} Calculation:
The output capacitor value will determine the amount of voltage ripple on the LED string. This voltage ripple, together with the dynamic resistance of the LED string will determine the current ripple through the LED string and this will cause 100Hz or 120Hz light flicker.
In this example the maximum allowed LED current ripple amplitude is set at 340mApp for a ripple percentage of 42%. The LED string uses 14 LEDs and has total dynamic resistance of 14Ω: V_{OUT} ripple = 0.34A*14 Ω = 4.76Vpp. The transformer secondary winding current can estimated having a low frequency ripple of double the line frequency and low frequency peak to peak amplitude of double the average output current. The output capacitor value can now be calculated:
where I_{OUT_PP} is 2x the average LED current, and V_{OUT_PP} is the allowed AC output voltage ripple and f is double line frequency. Calculating for 50Hz line frequency: C_{OUT }= 2*0.4/(4.76*2*π*100) = 267µF. For less LED current ripple, the C_{OUT} value needs to be increased. But when LED strings with higher dynamic resistance are used, the C_{OUT} value can be reduced.
Step 2. Transformer Design
Ideal turn’s ratio of primary to secondary windings can be expressed as
In the example:
V_{ro }= 125V, V_{O_max }= 47V, V_{f }= 0.7V, N_{P}/N_{S}= 2.62
Ideal turn’s ratio of secondary to auxiliary windings can be expressed as
In the example:
V_{O_max }= 47V, V_{DD_max }= 20V, N_{S}/N_{A }= 2.35
The maximum on time of the MOSFET ton_max can be expressed
in which f_{s_min} is the minimum switching frequency.
The duty ratio of the MOSFET D_{on} can be expressed
The primaryside inductance Lm can be derived as
Thus, L_{m} can be obtained after the minimum switching frequency f_{s_min} is determined.
In the example:
Set f_{s_min }= 54kHz,
V_{ro }= 125V, V_{ac_min_pk }= 127V,
Obtain t_{on_max }= 8.68μs and L_{m }= 899μH
The minimum number of turns for the transformer primary side to avoid the core saturation is given by:
where A_{e} is the crosssectional area of the core in m^{2}, and B_{max} is the maximum flux density in Gauss.
In the example:
I_{P_pk }= 1.23A, L_{m }= 899μH, EDR28 core is chosen and its A_{e} = 88m^{2}.
Set B_{max }= 2950 Gauss. Obtain N_{P_min} > 42.5 turns.
Now all the parameters of transformer are determined, including N_{P_min}, N_{P}/N_{S}, N_{S}/N_{A} and L_{m}.
N_{P} = 43T, N_{S }= 43/2.62 = 16.4T, choose 16T, N_{A }= 16/2.35 = 6.8T, choose 7T.
Step 3. Current Sense Resistor Determination
Current sense resistor R_{CS} can be determined as the following equation:
where K_{CC} is a reference in the controller.
In the example:
Actual N_{P}/N_{S }= 2.69, K_{CC }= 0.25, I_{O }= 0.4A, CTR_{TX1}= 0.9,
The current sense resistor R_{CS} will become (1/2)*2.69*(0.25/0.4)*0.9 = 0.79 Ω.
Step 4. Bridge Rectifier Determination
The maximum reverse voltage of the bridge rectifier V_{RRM_max} can be expressed as:
The maximum forward current of the bridge rectifier I_{BR_max} can be expressed as:
In the example:
I_{BR_max }= 22.12/90 = 0.25A
A 600V / 1A diode bridge will provide sufficient derating, including inrush current and voltage surge.
Step 5. MOSFET Determination
The maximum draintosource voltage stress of the MOSFET V_{DS_max} is given as:
in which V_{clamp} is the maximum voltage on the snubber and it must be higher than V_{ro}.
The maximum draintosource current stress of the MOSFET I_{DS_max} is given as:
In the example:
Set V_{clamp }= 160V
V_{DS_max }= 373+160 = 533V : for sufficient derating, choose at least 650V rated MOSFET.
I_{DS_max }= I_{P_pk }= 1.23A : MOSFET R_{dson} selection depends on thermal aspects. In this small T8 design, a 4A MOSFET with R_{dson }of 1.8Ω was selected, which can be used without heatsink.
Step 6. Output Diode and Auxiliary Diode Determination
The maximum reverse voltage stress of the output diode V_{Do_max} can be expressed as:
where V_{O_OVP} is the output over voltage level.
The maximum average forward current stress of the output diode I_{Do_max} can be expressed as:
In the example:
Set V_{O_OVP }= 61V
V_{Do_max }= 373/2.62+61 = 203V
I_{Do_max }= I_{O }= 0.4A
Diodes with higher current rating can be chosen for higher efficiency.
The maximum reverse voltage stress of the auxiliary diode V_{Da_max} can be expressed as:
where V_{DD_OVP} is the VDD over voltage level.
The maximum average forward current stress of the output diode I_{Da_max} can be expressed as:
in which I_{DD_max} is the maximum supply current for the controller.
In the example:
V_{DD_OVP }= 27V
V_{Da_max }= 373/(2.62*2.35)+27 = 87.8V
I_{Da_max }= I_{DD_max }= 5mA
Step 7. Minimum OnTime Setting
RT7302 limits a minimum ontime t_{on_min} for each switching cycle. The t_{on_min} is a function of the sampleandhold ZCD current I_{ZCD_SH} as following:
I_{ZCD_SH} can be expressed as:
Thus, R_{ZCD1} can be determined by:
In addition, the current flowing out of ZCD pin must be lower than 2.5mA (typ.). Thus, the R_{ZCD1} is also determined by:
In the example:
Set R_{ZCD1} =60kΩ
When V_{in} = 10V, t_{on_min} = 405p*60k*(2.62*2.35)/10 = 14.9μs
In general, longer t_{on_min} can slightly improve THDi. However, if t_{on_min} is too long, it will induce a current resonance at V_{in} zero crossing, worsening THDi. Thus, t_{on_min} can be properly defined according to the measured THDi.
Step 8. Output OverVoltage Protection Setting
Output OVP is achieved by sensing the knee voltage on the auxiliary winging. Thus, R_{ZCD1} and R_{ZCD2 }can be determined by the equation as:
In the example:
Set V_{O_OVP} = 61V
It can be calculated that R_{ZCD2} =7.9kΩ
Step 9. Propagation Delay Compensation Design
The V_{CS} deviation (ΔV_{CS}) caused by propagation delay effect can be derived as:
in which t_{d} is the delay period which includes the propagation delay of RT7302 and the turnoff transition of the main MOSFET. The sourcing current from CS pin of RT7302 I_{CS} can be expressed as:
where K_{PC} is a constant value in the controller. R_{PC} can be designed by:
t_{d} is estimated to be around 150ns
In the example:
R_{PC} = 150n*0.74*60k/(899μ*0.02)*(2.62*2.35) = 2.3kΩ
The delay period td is varied with the parasitic capacitance of MOSFET, the gate driving capability, and the propagation delay of the controller. Thus, t_{d} cannot be estimated accurately, and R_{PC} may need to be modified according to the measured output current. If the output current increases when V_{in} rises, R_{PC} should be increased. If the output current decreases with V_{in} rises, R_{PC} should be decreased.
Step 10. FeedForward Compensation Design (Only for RT7302)
The COMP voltage, V_{COMP}, can be derived from the following equations.
V_{MULT_pk} is the peak voltage on the MULT pin. Gm_{ramp} is the transconductance of the ramp generator, and its typical value is 2.5μA/V. C_{ramp} is the capacitance of the ramp generator, and its typical value is 6.5pF. When the converter operates at CRM, (t_{on }+ t_{off}) / t_{S}=1. V_{COMP_min} is recommended be within 1.2 ~ 1.5V, and R_{M2} is recommended to be within 30 ~ 60kΩ. Thus, the voltage divider resistors R_{M1} and R_{M2} can be determined according to the above parameters.
In the example:
t_{on_max }= 8.68μs.
Set V_{COMP_min }= 1.2V,
Obtain V_{MULT_pk }= 0.85V.
Set R_{M2 }= 43kΩ,
It can be calculated that R_{M1 }= 6.4MΩ.
4. Design Tool Explanation
The RT7302 design tool and RT7304 design tool can be used as quick way to determine the component values. The content is similar to the step by step design as described in Chapter 3. In the design tool, users input operation parameters into "Yellow Grid". According to the keyin parameter, the design tool will automatically generate the results in Pink Grid".
Table 1 below shows the entered data and calculation results for the 18W T8 reference design.
Table 1. Design tool data
5. Circuit Diagram of the Evaluation Board
The circuit diagram of the evaluation board is shown in Figure 4 below.
Figure 4. Schematic of the 18W T8 LED driver reference design
RV1 is added for line surge protection. LX2, CX1 and LX1 are added to reduce Line conducted EMI. L1 and LX3 are added to reduce radiated EMI.
The full BOM is shown in table 2 below.
Table 2. Full BOM of the 18W LED driver reference design
Item

Quantity

Reference

Part/Value

Type

Vendor

Remark

1

1

F1

T1.25A/300V

SS5F2P

Littlefuse


2

1

RV1

7N471K

TVS2P

Thinking


3

1

LX2

30mH

LRST14

Abliss

T12.7*7.92*4.9(μi=10000)

4

1

CX1

0.1μF

CFS12X12

Shiny Space


5

1

LX1

5mH

LDSD9X12

Mag. layers


6

1

BD1

1A/600V

DB1A

GW


7

1

C4

0.1μF/500V

CFS11X10

Murata


8

3

R6, R7, R9

2.2MΩ

0805

RALEC


9

1

R19

43kΩ

0603

RALEC


10

1

C6

22nF/50V

0603

Murata


11

1

C7

2.2μF/25V

0805

Murata


12

1

R22

0Ω

0603

RALEC


13

1

R8

10kΩ

1206

RALEC


14

1

R2

140kΩ

1206

RALEC


15

1

D2

FM4007

SOD123

Willas


16

1

C2

2.2nF/1kV

1206

Murata


17

1

R13

200Ω

0805

RALEC


18

1

D4

1N4148

SOD123

Willas


19

1

Q1

4A/650V

TO220

IPS

FTA04N65D

20

1

C9

100pF/1kV

1206

Murata


21

1

LX3

T3.5*3*1.4



King core

On Source pin of Q1

22

3

R15, R16, R17

2.21Ω

1206

RALEC


23

1

R14

2kΩ

603

RALEC


24

1

C10

470pF/1kV

1206

Murata


25

1

CY1

1000pF/250Vac

CAP10mm

Murata


26

1

R10

60kΩ

0603

RALEC


27

1

R18

8.06kΩ

0603

RALEC


28

1

C5

22pF

0603

Murata


29

1

D3

BAV21

SOD123

Willas


30

1

R11

82Ω

0603

RALEC


31

1

EC2

33μF/50V

CES5X11

Rubycon


32

1

T1

EDR28

EDR28

Abliss


33

1

U1

RT7302

SOP8

Richtek


34

1

D1

SF26

DO15

Willas


35

1

R1

33Ω

1206

RALEC


36

1

C1

220pF/1kV

1206

Murata


37

1

EC1

270μF/63V

CES10X25

Rubycon


38

1

L1

110μH

LRT9

Abliss

T9*5*3(μi=10000)

39

1

R4

200kΩ

1206

RALEC


Transformer design: The transformer design specification is shown in Figure 5.
Figure 5. Transformer specification
The primary side formed by sandwich structure is used to reduce the leakage inductance of the transformer, improving the efficiency and the output current regulation. To improve radiation EMI, the maximum voltage swing which is on pin 3 of the transformer, should be at the most inner side. To meet the safety standard, Triple wire at the secondary side is normally adopted for providing insulation.
6. Electrical Performance Measurements
Table 3 below shows the LED driver input and output parameters over the full mains voltage range.
Table 3. Performance measurements
Frequency

Vac [V]

Pin [Watt]

Vout [V]

Iout [mA]

Pout [Watt]

Eff. [%]

PF Value

THD

60Hz

90

21.54

45.75

405

18.53

86.02%

0.9960

6.37

60Hz

100

21.24

45.78

405

18.54

87.29%

0.9960

6.68

60Hz

110

21.03

45.80

404

18.50

87.98%

0.9954

7.03

60Hz

120

20.87

45.83

403

18.47

88.50%

0.9950

7.24

60Hz

132

20.73

45.86

402

18.44

88.93%

0.9944

7.53

50Hz

180

20.60

46.00

401

18.45

89.54%

0.9908

7.51

50Hz

200

20.60

46.07

400

18.43

89.46%

0.9886

7.02

50Hz

220

20.64

46.15

400

18.46

89.44%

0.9851

6.73

50Hz

230

20.69

46.23

400

18.49

89.38%

0.9832

6.82

50Hz

240

20.75

46.31

400

18.52

89.27%

0.9811

6.99

50Hz

264

20.90

46.44

400

18.58

88.88%

0.9738

7.86

Current regulation = 1.23%
ΔEfficiency = 3.52%
Maximum PF = 0.996
Minimum PF = 0.974
As can be seen, the current line regulation is excellent. Driver efficiency meets the target easily, and Power Factor and THDi are fully in line with regulations for lighting applications.
The figures in Table 4 show voltage and current waveforms during various operation conditions:
Table 4. Measured waveforms during various operation conditions
Startup



Vin = 90Vac: Tstart = 630msec

Vin = 264Vac: Tstart = 210msec

Input waveform



Vin = 90Vac

Vin = 264Vac

Output waveform



Vin = 90Vac

Vin = 264Vac

Harmonic content of input current: (IEC6100032)



Vin = 110Vac: passes Class C and D

Vin = 230Vac: passes Class C and D

Conduction EMI



Vin = 230VL

Vin = 230VN

The demo board passes conducted and radiated EMI at 120V and 230Vac

7. PCB Layout Information
The PCB layout of the reference design is shown in Figure 6 below. It is build from doublesided FR4 material and uses a narrow formfactor to make it suitable to fit into narrow T8 enclosures.
To minimize EMI, current loops of the gate drive, snubber circuit, output diode and main MOSFET switching loop should be kept as small as possible. Ground of the IC, sense resistor, aux winding and Ycapacitor should all go to one central ground point on the input capacitor ground. Capacitors on the IC COMP pin, ZCD pin and MULT pin should be as close as possible to the IC.
Top silk (Component location)
Top trace
Bottom trace
Figure 6. PCB Layout
8. Summary
With the help of this step by step design guide and the RT7302 design tool, the user is able to quickly design a LED driver that fulfills the requirements for high performance offline LED drivers. The absence of secondary side sensing greatly simplifies the mechanical design, and allows small formfactor PCB design.
When all guidelines are followed, the design should fulfill EMI and pass the surge tests. Although this reference design is for an 18W LED driver, RT7302 can be used in a wide range of LED driver designs, ranging from 8W ~ 60W.